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DSP-Based Electromechanical Motion Control (Power Electronics and Applications Series)

This implies that the instantaneous mains voltage is greater than the capacitor voltage only for very short periods of time charging time of capacitor. During this short period the capacitor must charge fully. Therefore, large pulses of current are drawn from the line over a very short time. This is true for most of the rectified AC signals with capacitive filtering. Depending on the loading on the utility, the current shape will be different.

The load can be linear, such as in resistive heaters, or it can be nonlinear, such as in an advanced air conditioner with switching power supply. As mentioned in the first section of the chapter, mostly these switch- ing power supplies have diode rectifiers on their front ends. The current waveforms are no longer sinusoidal for them, and thus the definition of power factor is changed for them. Although they are not enforced in the United States, they have attracted a lot of attention in the power electronics industry. In most appli- cations it is not difficult to meet these standards; however, the most eco- nomic choices are still being developed.

IEEE and international harmonic standards can be grouped into three main categories: 1. The IEC series deals with electromagnetic compliance. IEC not only deals with individual equipment but also sets limits for the whole system installation. Both single-phase and three- phase harmonic limits are addressed in this section of the regulation. The philosophy behind this standard is to prevent harmonic currents from traveling back to the power system and affecting other customers. Input PFC means a system in which the power factor correction circuit is placed at the input of the offending network, such as switched-mode power supply SMPS.

When loads are highly reactive, for example, bal- lasts, the PFC circuit is placed between this kind of load and the power supply. This circuit is known as output PFC. Depending on the components used to develop the PFC circuits, they can also be classified in two ways. When simple reactive components, such as inductors and capacitors, are used to correct the displacement between line voltage and line current, the process is known as passive PFC.

Both of them have their relative merits and demerits. Active PFC circuits are generally used to compensate for distortion of the mains current. These circuits are mostly switched-mode power sup- ply topologies. This means that they are more complicated than passive circuits. With the advancements in integrated circuit IC technology, they are becoming simpler, more compact, and also less costly. From the operating frequency point of view, they can be either slow switching topologies or high switching topologies. Next, we discuss high switching topologies in detail. They are supported by control ICs from most of the major manufactur- ers.

High frequency circuits offer many advantages over passive and low frequency techniques.

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Such advantages are low weight, accurate voltage control, low line harmonic distortion, wide operating voltage range, and easy design. Their performance characteristics, block diagrams, suitability to the application, and relative advantages and disadvantages are discussed in the following subsections. The pre-regulators are designed to draw an input current, which varies in direct proportion to the instantaneous input voltage. The control circuits control the root mean square RMS current drawn from the line. Though the circuits are more complex, power factor of almost unity is achievable with these high frequency topologies.

All the topologies can operate in both discontinuous conduction mode DCM as well as in continuous conduction mode CCM mode. Power factor correction is therefore not achieved when the instantaneous input voltage is lower than the required output voltage [1]. This topology, on its own, is not suitable for PFC. Improved performance is obtained if cascaded with a boost circuit. The buck topology has a switch in series with the mains or supply line. Thus, the input current is switched directly, which in turn can generate EMI problems.

The drive circuit for the switch is more complex than that for the boost circuit. The advantages with buck can be summarized as follows: less stress on the bulk capacitor, voltage rating of the switch is much less than the boost one for the same power rating of the converter , and inrush current protection is inherent [2].

It must be noted that if the peak input voltage exceeds the controlled capacitor voltage then power factor correction is not achieved Table 3. The input current from the mains is not chopped directly in this topology, because the inductor is placed in series with the source. And this helps with EMI, but care must be taken with the high fre- quency current that is drawn from the line. Fortunately, the series choke at the input of the PFC circuit helps to absorb some of the line transients.

The voltage across the switch is only the output voltage—low voltage source on the switch. This method does not offer inrush current limiting as the input inductor is comparatively small, so this must be accomplished by other means. The advantages of the boost converter can be summarized as follows: good EMI performance, voltage rating of the switch is a bit less than other topologies, it absorbs line transients, and easy drive circuits are required.

On the negative side, it offers no PFC control when input voltage is higher than the output voltage, and it cannot limit the inrush currents Figures 3. So we can say that input current follows the input voltage [5], [6]. At the same time the output voltage is allowed to vary independently in a specified range Table 3. But this topology too has limi- tations.

One of them is that the polarity of its output voltage is reversed. This means that the input and output should have separate grounds. And the other limitation is that the drive circuit for this topology is floating. Here also we have two inductors, one at input and the other at output.

It also depends on the discharge time of the input inductor. Here the advan- tage is that the output voltage can vary independently in a specified limit as in the buck-boost topology Figures 3. As mentioned ear- lier, to achieve natural power factor correction [8], it is desirable to operate the input inductor in DCM Figure 3. In other words, neither of them changes its state of energy. From the state equations for this converter, it can be said that the voltage-current relationship depends on the switch- ing frequency and duty ratio Table 3.

This means that the state of energy of each energy storage element is not independent. The input inductor inherently makes this topology suitable when isolation is necessary, because this inductor can be utilized as a magnetizing inductance of the transformer. It also provides the feature of completely demagnetizing the transformer core. This topology is better suited for higher power applications, such as to W applications, than the flyback converter for the same applications.

From the topology shown in Figure 3. This can be achieved by duty ratio as well as transformer turns ratio [10]. Having the switch in series with the input provides the feature of overload protection. At the same time, it also chops the input current and generates EMI, which is to be filtered explicitly. The problem with flyback is the difficulty in programming the input current half sine wave when using current mode control CMC.

This is because CMC controls the peak inductor current, which is the input current in the boost but is not in the flyback. The input and induc- tor currents vary quite a bit with input voltage in the flyback topology. The solution to this problem lies in average current mode control, but the control circuit is more complex.

The advantages can be summarized as follows: output can vary independently, it provides current protection, and it can provide input and output isolation. Problems can be summa- rized as follows: it requires a large filter for EMI, and the switch voltage rating is the sum of peak input voltage and output voltage Figures 3. But it is absolutely necessary to have third winding to discharge magnetizing current. Even though third winding is provided, the diode rectifier at the input prevents the negative current and one can easily conclude that this topology is not suitable for PFC in any case.

This topology is much better suited for high power post regulators Figures 3. A power electronic interface is absolutely necessary for running it optimally. C-dump converter topol- ogy is one of the popular topologies to drive a switched reluctance motor. As this topology contains the maximum number of passive elements and complexity, it is chosen to verify the PFC function for advanced drives, as a worst-case scenario.

Figure 3. As mentioned in Section 3. This will deteriorate both power factor and overall system performance. Figures 3. From the supply voltage and current waveforms shown, it is clear that extensive distortion occurs on the supply side. As mentioned in the previous sections, the boost converter is the best- suited topology for active power factor correction. From the waveforms shown in Figure 3. No current peaks are visible in the supply current waveform.

Explanation of the detailed operation of the boost converter can be found in the references. Here, the boost converter switch is controlled, keeping output voltage of the converter in mind, and it has nothing to do with controlling the switches of the main SRM drive circuit. But during dynamic conditions it should be observed that the overall system is not going to be unstable. Sometimes all the switches in a system are synchronized to avoid this problem. In this chapter, the reasons for employing power factor correction tech- niques and various methods of PFCs and their effectiveness are discussed.

Simulation results have been provided to justify the theoretical analysis. International and IEEE standards impose limits on harmonic voltage and current. Compliance with the IEC standards has been studied with computer simulations. The effectiveness of active PFC is normally not a problem, but the cost involved in the additional power electronics circuit could be a major obstacle to acceptance.

Although using these passive PFC methods complies with standards found satisfactory, the problems of EMI, electromagnetic compatibility EMC , and the size of the passive elements involved in them are to be justified. Wu, H. Yuan, J. Zhang, and W. Novel single phase cur- rent source buck PFC with delta modulation control strategy. In Sixth inter- national conference on power electronics and variable speed drives.

Hirachi, K. Improved control strategy on buck-type PFC converters.

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Electronic Letters 31 12 — Ping L. Yu, K. Yong, Z. Hui, and J. Analysis of single-phase boost power factor correction converter. Gusseme, D. Sype, A. Bossche, and J. Nishida, Y. Motegi, and A. A single-phase buck-boost AC-DC converter with high quality input and output waveforms.

Matsui, K. Yarmanoto, T. Kishi, M. Hasegawa, H. Mori, and F. Comparison of various buck-boost converters and their application to PFC. In IEEE 28th annual conference of the industrial electronics society. Tseng, C. A novel ZVT Cuk power factor corrector. Buso, S. Spiazzi, and D. Cruz, M. Desouza, and I. Zeta converter with high power factor operating in CCM. Kim, S. A parallel connected single-phase power factor correction approach with improved efficiency.

In Applied power electronics conference and exposition. An SMPS offers three main advantages over a conventional linear power supply: high efficiency and less heat generation, tighter regu- lation, and small size and weight. Conventional linear power suppli- ers are inefficient because they regulate by dumping the excess power into heat. Another key benefit of SMPSs is their ability to closely regulate the output voltage. Switched-mode supplies regulate continuously and follow load changes almost immediately. In addition, switchers have the unique ability to maintain the correct output under low input conditions.

In fact, switch- ers can actually produce an output voltage that is higher than the DC voltage applied to the input. A final advantage of switchers is their rela- tively small size and weight.


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Because switches operate at high frequen- cies, the parts are smaller than those needed for a conventional 60 Hz power supply of small power rating. The transformers, capacitors, and coils are also physically smaller and lighter [1]-[5]. There are many important applications for switched mode power supplies.

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There are two kinds of general control methods for switched mode DC-DC power supplies. One is voltage mode control. The other one is the current mode control. There are three current control methods: tolerance band control, constant off time control, and constant frequency with turn on at clock time [16]—[19]. They are shown in Figures 4. Compared to the line power supply, this method does have advantages for using the high switching frequency [10], [11]. Simplification means the process of synthesizing converters with reduced components, smaller size, and lower weight or cost.

System objec- tives can include minimum cost, maximum efficiency, high reliability, low switch stress and power packing, wide conversion range, PFC and output regulation, inverter PFC, and better performance. Integrated converters own at least two con- verter sets, and the basic converters, such as buck and boost, only own one converter set. In general, every converter set has some relationship with the other.

It can be seen from the configuration of integrated converters that the integrated converter not only owns all discrete functions of every converter set, but also has a simplification process based on the system integration. Care should be taken that each discrete converter set can be a sub-integrated converter or basic converter.

This kind of sub-inte- grated converter is based on the integration of every sub-building block. The integrated converter is based on the integration of a sub-integrated converter [5]. It means that the integrated converter is based on the sys- tem integration.


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  6. The application field for conventional switched- mode power supplies is also suitable for integrated switched-mode power supplies. The traditional cascade power converter is replaced by an inte- grated power converter. In this way, the new power supplies can have higher efficiency or higher density or smaller size or better output regula- tion than traditional switched-mode power supplies [10], [11]. Integrated converters can not only implement the system functions similar to the discrete converters without integration, but they also have other special functions.

    Integrated converters can be classified into four types. Each of them will be intro- duced in the following chapters. The control methods for the conventional switched-mode power supplies are also suitable for the integrated converters, which have more complicated transfer functions. New, flexible control methods are also needed. Blaabjerg, F, A. Consoli, J. Ferreira, and J. The future of electric power processing and conversion.

    Bose, B. Power Electronics and Variable Frequency Drives. Emadi, A. Multi-converter power electronic systems: definition and application. Modeling and analysis of multiconverter DC power elec- tronics systems using the generalized state-space averaging method. Ericsen, T. Power electronics building block—one system approach to power electronics.

    In Power engineering summer meeting. Aguilar, C. Canales, J. Arau, J. Sebastian, and J. Bekiarov S. B, and A. Uninterruptible power supplies: classi- fication, operation, dynamics, and control. Lee, F. The state of art power electronics technologies and future trends. In Power engineering society summer meeting.

    Krishnan R. Topologies for uninterruptible power supplies. Liu, X. Haque, J. Wang, and G. Packaging of integrated power electronics modules using flip-chip technology. Madigan, M. Erichson, and E. Integrated high-quality rectifier-regulators. Switching Power Supply Design. New York: McGraw- Hill. Shepard, J. Power electronics future. Vanwyk, J. Power electronics technology at the draw of the new millenium status and future.

    Wilson, T. The evolution of power electronics. Banerjee, S. Nonlinear phenomena in power electron- ics: attractors, bifurcations, chaos, and nonlinear control. Bartoli, M. Reatti, and M. Open loop small signal control to output transfer function of PWM buck converter for CCM: model- ing and measurements. Ross, J. The Essence of power electronics. New York: Prentice Hall.

    Stemier, P. Power electronics building block—a platform based approach to power electronics.

    Ou electrical syllabus

    In Power engineering society general meeting. Zhong, N. Emadi, J. Mahadavi, and A. In International telecommunications energy conference, Montreal, Canada. IEEE, Sept. Murali, V. In Power electronics specialists conference. Willers, M. Egan, J. Murphy, and S. Both of them share one active switch. The output of the boost converter will be used as the input source for the second converter. When the switch is on, the boost inductor is charged. The second converter will also work in the switch-on state. When the switch is off, the inductor is discharged.

    The sum of the input source energy and the inductor storage energy will be transferred to the boost output [1]—[7]. The boost converter consists of inductance Lin, diode D1, diode D2, switch Q, capacitor C1, and input source vs. The flyback converter consists of the capacitor C1, switch Q, transformer, diode D3, output capacitor Cout, and load resistor R where the flyback con- verter and boost converter share a common switch Q. The input source Vc1 for the flyback converter is from the output of the boost converter.

    The boost converter consists of inductance Lin, diode D1, diode D2, switch Q2, capacitor C1, and input source vs. The double-ended flyback converter consists of the capacitor C1, switch Q1, switch Q2, transformer, diode D3, output capacitor Cout, and load resistor R where double-ended flyback converter and boost converter share a common switch Q2.

    The input source Vc1 for the double-ended fly- back converter is from the output of the boost converter. The parallel flyback converter consists of the capacitor C1, switch Q1, switch Q2, transformers, diode D3, diode D4, output capacitor Cout, and load resistor R where the parallel flyback converter and boost converter share a common switch Q1. The input source Vc1 for the parallel flyback converter is from the output of the boost converter.

    The forward converter consists of capacitor C1, switch Q, transformer, diode D3, diode D4, output inductance Lout, output capacitor Cout, and load resistor R where the forward converter and boost converter share a com- mon switch Q. The input source Vc1 for the forward converter is from the output of the boost converter. Lout, output capacitor Cout, and load resistor R, where the double-ended forward converter and boost converter share a common switch Q2.

    The input source Vc1 for the double-ended forward converter is from the out- put of the boost converter. The boost converter consists of inductance Lin, diode D2, switch Q1, capacitor C1, capacitor C2, and input source vs. The input source Vc1 and Vc2 for the series parallel forward con- verter is from the output of the boost converter. The parallel forward converter consists of the capacitor C1, switch Q1, switch Q2, transformers, diode D3, diode D4, diode D5, diode D6, output inductance Lout, output capacitor Cout, and load resistor R, where the paral- lel forward converter and boost converter share a common switch Q1.

    The input source Vc1 for the parallel forward converter is from the output of the boost converter. The input source for the full-bridge Vc1 is from the output of the boost converter. The half-bridge converter consists of the capacitor C1, capacitor C2, switch Q1, switch Q2, transformer, diode D2, diode D3, diode D4, output inductance Lout, output capacitor Cout, and load resistor R, where the half- bridge converter and boost converter share a common switch Q2.

    The input source for the half-bridge Vc is from the output of the boost converter. The push-pull converter consists of the capacitor C1, switch Q1, switch Q2, transformer, diode D3, diode D4, output inductance Lout, output capaci- tor Cout, and load resistor R, where the push-pull converter and boost con- verter share a common switch Q1. The input source for the push-pull Vc1 is from the output of the boost converter [7]. This converter uses two inductances. The steady state average voltage across C1 is always equal to the input voltage Vin.

    The Sepic converter has a special advantage in high power factor pre- regulation applications. If two inductors are coupled, ripple current steer- ing can be achieved. The only difference is the input diode in series with the input inductor. This diode prevents the negative current flow in the line. Therefore, it is possible to achieve those operating modes that are not possible with the isolated SEPIC converter [2]-[3].

    For the buck converter, it can be non-isolated or isolated. With what Dr.

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    The correct use of this topology is as a regulator in a standard power supply, whether off-line or from a DC source. It is also suitable for invert- ing on-board regulators, complementing the boost-buck cascade. To null output ripple, the two inductors may be coupled. Winding them on the same core with a gap to reduce the coupling coefficient to 0. The procedure yields a control function with a single RHP zero, which cannot be damped out.

    The lack of output ripple eliminates high frequency heating of the speaker magnet, leaving a clean output waveform. If an isolation transformer is used in this converter, two inductors and the transformer can be integrated. This will result in zero input and output ripple. In this case, the converter is ideal for the practical model of a DC transformer. The only difference is that the input diode is in series with the input induc- tor [4]. This input diode prevents the negative current flow in the line. Johnston, M. In Applied power elec- tronics conference and exposition proceedings.

    Albrecht, J. Yong, and W. Boost-buck push-pull converter for very wide input range single stage power conversion. Degusseme, K. Vande, A. Vanden, and J. Proceedings of the 34th annual IEEE power electronics specialists conference. Nie, Z. Ferdowsi, and A. Boost integrated push-pull recti- fier with power factor correction and output voltage regulation using a new digital control technique.

    Proceedings of the IEEE international telecommu- nications energy conference. The output of the buck converter will be used as the input source for the second converter. When the switch is on, input source energy will transfer to the buck output. The second converter will also work in the switch on state. When the switch is off, input source energy for the buck converter is disconnected from the buck output [1]-[5]. The buck converter consists of inductance Lin, diode D1, diode D3, switch Q, capacitor C1, and input source vs.

    The flyback converter consists of the capacitor C1, switch Q, trans- former, and diode D2, diode D3, diode D4, output capacitor Cout, and load resistor R where the flyback converter and buck converter share a com- mon switch Q. The input source Vc1 for the flyback converter is from the output of the buck converter [1]. The double-ended flyback converter consists of the capacitor C1, switch Q1, and switch Q2, transformer, diode D2, diode D3, diode D4, out- put capacitor Cout, and load resistor R where the double-ended flyback converter and buck converter share a common switch Q2.

    The input source Vc1 for the double-ended flyback converter is from the output of the buck converter [1]. The parallel flyback converter consists of the capacitor C1, switch Q1, switch Q2 transformers, diode D2, diode D3, diode D4, diode D5, output capacitor Cout, and load resistor R, where the parallel flyback converter and buck converter share a common switch Q1.

    The input source Vc1 for the parallel flyback converter is from the output of the buck converter. The double-ended forward converter consists of the capacitor C1, switch Q1, switch Q2, transformer, diode D3, diode D4, diode D5, output inductance Lout, output capacitor Cout, and load resistor R, where the dou- ble-ended forward converter and buck converter share a common switch Q2. The input source Vc1 for the double-ended forward converter is from the output of the buck converter [1].

    The buck converter consists of inductance Lin, diode D1, diode D3, switch Q1, capacitor C1, and input source vs. The series parallel forward converter consists of the capacitor C1, capacitor C2, switch Q1, transformer, diode D1, diode D2, diode D3, diode D4, diode D5, diode D6, diode D7, output inductor Lout, output capacitor Cout, and load resistor R. The series parallel forward converter and buck converter share a common switch Q1. The parallel forward converter consists of the capacitor C1, switch Q1, switch Q2, transformers, diode D3, diode D4, diode D5, diode D6, diode D7, the output capacitor Cout, output inductance Lout, and load resistor R, where the parallel forward converter and buck converter share a common switch Q1.

    The input source Vc1 for the parallel forward converter is from the out- put of the buck converter [1]. The buck converter consists of inductance Lin, diode D1, diode D3, switch Q2, capacitor C1, and input source vs. The full-bridge converter consists of the capacitor C1, switch Q1, switch Q2, switch Q3, switch Q4, transformer, diode D3, diode D4, diode D5, diode D6, output inductance Lout, output capacitor Cout, and load resistor R, where the full-bridge converter and buck converter share a common switch Q2.

    The input source for the full-bridge Vc1 is from the output of the buck converter. The half-bridge converter consists of the capacitor C1, capacitor C2, switch Q1, switch Q2, transformer, diode D2, diode D3, diode D4, output inductance Lout, output capacitor Cout, and load resistor R, where the half- bridge converter and buck converter share a common switch Q2. The input source for the half-bridge Vc is from the output of the buck converter. The push-pull converter consists of the capacitor C1, switch Q1, switch Q2, transformer, diode D3, diode D4, diode D5, output inductance Lout, out- put capacitor Cout, and load resistor R, where the push-pull converter and buck converter share a common switch Q1.

    The input source for the push- pull Vc1 is from the output of the buck converter [1]. Integrated switched-mode power supplies. PhD dissertation, Illinois Institute of Technology. Bryant, B. Chen, J. Analysis and design of SEPIC converter in boundary conduction mode for universal-line power factor correction appli- cations. Kursun, V. Narendra, V. De, and E. Analysis of buck converter for on-chip integration with a dual supply voltage micropro- cessor. We take two basic paral- lel converters as an example. They share one active switch and one input source voltage. When the integrated converter works in the fixed switching frequency and duty ratio with the open control loop, the output voltage can be regu- lated by designing the suitable converter parameters.

    When it works in the closed control loop, possible control methods can be as follows: a by studying converter characteristics, only regulate one of them; b alter- nately regulate two output voltages. Multi-output voltages can provide custom outputs different output voltages and quality. On the other hand, it increases the density of the converter. The con- verters share a common switch Q and input voltage. One buck converter consists of these common components: switch Q, diode D1, diode D2, inductor Lin1, output capacitor Cout1, and resistor R with output voltage vo1.

    The other buck converter consists of the following common components: switch Q, diode D3, inductor Lin2, output capacitor Cout2, and resistor R with output voltage vo2. The buck con- verter consists of these common components: switch Q, diode D1, diode D2, inductor Lin1, output capacitor Cout1, and resistor R with output voltage vo1.

    The buck consists of common components switch Q, diode D1, diode D2, inductor Lin1, output capacitor Cout1, and resistor R with output voltage vo1. One buck-boost converter consists of common components switch Q, diode D1, diode D2, output capacitor Cout1, inductor Lin1, and resistor R with output voltage vo1. The zeta converter consists of com- mon components switch Q, diode D1, diode D2, capacitor Cout1, induc- tor Lin1, inductor Lin2, and output resistor R with output voltage vo1.

    The buck-boost converter consists of common components switch Q, diode D3, diode D4, capacitor Cout2, inductor Lin2, and output resistor R with output voltage vo2. For the zeta plus zeta double system, the converters share a common switch Q and input voltage. One zeta converter consists of common components diode D1, diode D2, capacitor C1, capacitor Cout1, inductor Lin1, inductor Lin2, and output resistor R with output voltage vo1.

    The other zeta con- verter consists of common components diode D3, diode D4, capacitor C2, capacitor Cout2, inductor Lin1, inductor Lin2, and output resistor R with out- put voltage vo2. One boost converter consists of common components diode D1, capacitor Cout1, and resistor R with out- put voltage vo1. The other boost converter consists of common compo- nents diode D2, capacitor Cout2, and resistor R with output voltage vo2.

    The boost converter consists of common components, diode D1, capacitor Cout1, and resistor R with output voltage vo1. The flyback converter consists of common components diode D2, capacitor Cout2, and resistor R with output voltage vo2.

    Digital Signal Processing & Power Electronics Realistic Interview, or Viva Voce

    The forward converter consists of com- ponents diode D1, Capacitor C2, inductor L1, capacitor Cout1, and resistor with output voltage vo1. The boost converter consists of common compo- nents diode D3, capacitor Cout2, and resistor R with output voltage vo2. The forward converter consists of common components diode D1, diode D2, transformer, inductor Lout, capacitor Cout1, and resistor R1 with output volt- age vo1.

    In International telecommunications energy conference. In Power electronics specialist conference. In Industrial electronics, control and instrumentation confer- ence. Boost-buck push-pull con- verter for very wide input range single stage power conversion. Prado, R. A high-power-factor electronic ballast using a flyback push-pull integrated converter.

    Wang, C. A novel single-stage full-bridge buck-boost inverter. It is nearly always a good approximation to assume that the magnitude of the ripple is much smaller than the DC component [1]—[4]. This method greatly simplifies the analysis of the converter waveforms. In steady state the initial and final value of the inductor current are equal. In steady state the initial and final values of the capacitor voltages are equal.

    In mode 5, the DC voltage conversion ratio, m2 , can be obtained as Equation 9. An analytical technique for the analysis of switching dc-dc converters. In IEEE international symposium on circuit and systems. Middlebrook, R. A general approach at modeling switching converter power stages. In proceedings of the ieee power electronic specialists conference.

    In Conference on industrial electronics, control and instrumenta- tion. Cuk, S. A general unified approach to mod- eling switching DC-DC converters in discontinuous conduction mode. First, average vari- able value will be thought of as the sum of steady state value and small perturbation variable value [4], [5]. Second, higher order small perturba- tion will be ignored in the final model and calculation. Finally, the DC transfer model is used for the small signal model.

    The small signal model for the buck integrated forward converter in a different model will be derived in the following section [2]. The buck converter consists of inductor Lin, diode D1, diode D3, switch Q, capacitor C1, and input source vs. The forward converter con- sists of capacitor C1, switch Q, transformer, diode D3, diode D4, diode D5, output capacitor Cout, output inductor Lout, and load resistor R.

    In the buck integrated forward converter, the buck converter and forward converter share a common switch Q. In order to reset the transformer winding during every period, the maximum duty for this buck integrated forward converter is less than 0. When switch Q is on, capacitor C1 discharges through the transformer, switch Q, and diode D2 for supplying, together with current iLin, the for- ward converter.

    At the same time, input voltage charges the capacitor C1 by inductor Lin, capacitor C1, diode D3, and switch Q. When the switch is off, the current on the inductor Lin can be continuous and discontinues. So does the current on the output inductor Lout. Two kinds of DCM modes are classified by the different times of inductors reaching the zero current after switch is off Figures Two kinds of DCM modes are classified by the different time of induc- tors reaching the zero current after switch is off Figures In order to reset the forward converter in every cycle, the maximum duty ratio for the buck integrated forward converter is 0.

    When the turn ratio for the forward converter transformer is great than 2, the overall voltage ratio is less than that of the forward converter. Transfer function for different modes is analyzed in the following text. Figure This chapter compares synchronous rectifiers to Schottky diode types and illustrates some applications. The next generation of portable products, such as personal communi- cators and digital assistants, will have to provide at least 12 hours of oper- ation between battery charges.

    Most of the progress toward this hour goal must come from radio frequency RF , computer, and battery technol- ogy, because power-supply performance is approaching a limit. Still, the power supply must squeeze a battery for all it is worth. A key element in this task, especially at the low output voltages that future microprocessor and memory chips will need, is the synchronous rectifier. A synchronous rectifier is an electronic switch that improves power- conversion efficiency by placing a low-resistance conduction path across the diode rectifier in a switched-mode regulator [1]. Metal-oxide— semiconductor field-effect transistors MOSFETs usually serve this pur- pose, but bipolar transistors and other semiconductor switches can be considered for typical applications.

    The forward-voltage drop across a switched-mode rectifier is in series with the output voltage, so losses in this rectifier determine effi- ciency almost entirely [2], [3]. As supply voltage decreases, the design of rectifiers requires more attention, because the forward-voltage drop constitutes an increasing fraction of the output voltage. The race to new voltage levels proceeds in jumps, as each major chip manufacturer brings successive fabrication processes on line. Currently, research indicates a VCC of 1. Even designers of desktop PCs and workstations are turning to synchronous rectification as their power requirements increase and new ICs ease their implementation.

    However, the use of NMOS devices in a high-side configuration com- plicates the design. An auxiliary supply, a bootstrap circuit, or a charge pump must bring the gate voltage above the source node input voltage. This section deals with application of the synchro- nous rectification technique to basic switching regulator topologies, for example, buck, boost, and buck-boost [4], [5]. The performance of these con- verters is examined under this operating condition. The inductor and capacitor act as a lowpass filter to restore a nearly constant output voltage.

    Synchronous rectification increases the efficiency of a buck converter by replacing the Schottky diode with a low-side NMOS Figure This synchronous switch operates in the third quadrant, because the current flows from the source to the drain, which results in a negative bias across the switch. A positive voltage at the gate of the device still enhances the channel.

    Under light loads, the control block not shown in Figure The dead time eliminates the possibility of a destructive shoot-through condition; for example, both MOSFETs conduct simultane- ously. Standard designs use the same method to delay the turn-on of the main switch. This stored charge must sweep out to allow the body diode to recover its forward-blocking characteristic.

    The Schottky diode can have a lower current rating than the one that the standard buck regulator uses. This is because the diode conducts only during the dead times, which lowers the RMS current. It can be said that the diode conducts for all the time the main switch is off. All this time, the forward voltage drop occurs and that causes significant power loss Figure This will result in low power loss, and thus improvement in the efficiency.

    Even at very low voltages, rectifier loss is significant. For step-down regulators with a 3. The losses are not as bad at lower input voltages, because the rectifier has a lower duty cycle and thus a shorter conduction time. For an input voltage of 7. This figure also shows that, as output voltage decreases, the synchronous recti- fier provides even larger gains in efficiency. The operation of the synchronous rectified boost circuit is the same as that for the conventional, and the differences are the same as those men- tioned for the buck converter in the previous section.

    Similar to the boost topology, the inverting topology connects the synchronous rectifier in series with the output rather than to ground. Although these features would appear to favor current-mode con- trol in applications that require a fast dynamic response, this control method has some disadvantages. For example, it tends to be sensitive to noise in the control loop. Finally, the controller uses a current-sensing resistor in series with the output inductor. Voltage-mode control is attractive for low-voltage buck converters, because it involves a single control loop, exhibits good noise immunity, and allows a wide range for the PWM duty cycle ratio.

    Also, voltage- mode converters do not require a resistor for sensing current. However, the transfer function of standard voltage-mode buck converters that use Schottky diodes changes from no load to full load, making it difficult to achieve fast response to large dynamic loads. In the standard buck converter it has been observed that the light-load transfer function exhibits no double pole at the LC filter frequency that is characteristic of the full-load transfer function.

    This difference occurs because the Schottky diode in the standard buck configuration allows inductor current to flow only in one direction. Note that current-mode converters do not exhibit this behavior. The transfer function of a current-mode converter changes only slightly from discontinuous operation to continuous operation. The current-mode con- troller has two loops. The purpose of the inner, or current, loop is to divide the high-Q double pole of the LC filter into two single, well-separated, low-Q poles.

    Discontinuous operation exhibits a single low-frequency pole. Due to the popularity of current-mode control over voltage-mode control for synchronous rectifiers, basic current-mode control methods are briefly discussed in next section. They implement different control schemes, although there are three basic prin- ciples that reside at the heart of any control scheme. First, it is possible to continue to hold the synchronous switch on until the beginning of the next cycle, allowing the inductor to reverse. Second, you can completely dis- able the synchronous rectifier at light loads.

    Each approach involves a trade-off in different areas. In the past, the option that designers widely used was holding the inductor switch on until the beginning of the next cycle, which requires driving the MOSFET gates with complementary waveforms. This approach produces lower noise and allows a simple control scheme: the gate-drive signal is simply an inverted, opposite-phase version of the drive signal for the high-side switch. Noise is lower for two reasons, both of which relate to the continuous inductor current. First, the absence of pulse skip- ping ensures a constant switching frequency, regardless of load.

    A con- stant, fundamental switching frequency ensures that output ripple and EMI at the harmonic frequencies will not cause havoc in the intermediate frequency IF bands of an audio or radio system. Second, this approach eliminates the dead time during which a resonant tank circuit comprising the inductor and stray capacitance at the switching node can introduce ringing.

    The drawback of letting the inductor current reverse is that the syn- chronous rectifier pulls current from the output. The circuit replaces this lost output energy during the next half cycle. However, at the beginning of the cycle when the high-side switch turns on, the circuit transfers the inductor energy stored during the earlier current reversal to the input- bypass capacitor.

    This action resembles perpetual motion, in which energy shuttles between the input and output capacitors. Unfortunately, friction spoils all perpetual-motion schemes. As energy shuttles back and forth, the circuit dissipates power in all its tiny parasitic resistances and switching inef- ficiencies.

    Thus, additional energy is necessary to maintain the shuttling action. The most obvious consequence is a high no-load supply current of typically 5 mA for the 2. The second option, turning off the synchronous rectifier entirely at light loads, offers simplicity and low quiescent supply current. You usually implement this method in conjunction with a pulse-skipping operation, governed by a light-load pulse-frequency modulation PFM control scheme. Whenever the circuit goes into its light-load pulse-skipping mode, the circuit disables the synchronous rectifier that lets an accompanying parallel Schottky diode do all the work.

    Disabling the synchronous recti- fier prevents the reversal of inductor current, and the problem of shuttling energy back and forth does not arise. This method provides the best light-load efficiency, because the synchronous rectifier does its job without allowing the inductor current to reverse.

    During the switching dead time, current flows through the parasitic diode. A control scheme can shift the synchronous rectifier operation from the complementary-drive option to the off-at-zero option. The control circuit should employ PWM for heavy loads and automatically switch to a low quiescent current pulse-skipping mode for light loads. Such control comes in handy for computers with built-in radios. When the radio is not in use and the host system goes from run mode to suspend mode, the power supply automatically assumes its light-load pulse-skipping mode to save power.

    If the RF transceiver is turned on, a logic signal forces the supply to a low-noise mode that maintains quiet operation, regardless of output load. These advantages include improved efficiency, higher switching frequency, reduced EMI, and simplified thermal design. However, in contrast to a converter with discrete MOSFETs, an inte- grated design takes advantage of matched-silicon parameters.

    Worst- case analysis is less severe, because parameters such as gate charge and threshold tend to track with process variations and operating conditions. Integrated power devices also reduce parasitic inductances from the critical high-speed connections. These performance improvements let one build a converter that reduces dead time to less than 20 nsec, switches with rise and fall times lower than 10 nsec, and operates at frequencies higher than 1 MHz.

    When discrete MOSFETs are used, which vendors fabricate using vertical technologies, the substrate is at drain potential. Thus, conductive cooling requires large printed circuit PC traces. In contrast, the substrate and the tab of the integrated package are at ground potential. Therefore, heat can transfer directly from the power switches, through the tab, and then to the ground plane. Why not integrate all the required silicon for the synchronous regulator into a single IC?

    This level of integration is achievable but involves trade- offs. An IC that integrates the PWM controller, power switches, and drive and synchronous control has greater die size and pinout. Power IC packages with the required pin count and thermal capabilities are expensive.

    For example, it is possible to reduce gate loss by using a gate drive of 5 V as for logic-level MOSFETs instead of the input battery voltage. This approach adds complexity in the form of a bypass switch for the initial power-up. To prevent switching overlap of the main switch and synchronous rectifier MOSFETs that might cause destructive cross-conduction cur- rents, most switching regulators include a dead-time delay.

    The synchro- nous rectifier MOSFET contains an integral, parasitic body diode that can act as a clamp and catches the negative inductor voltage swing during this dead time. Therefore, designers interested in squeezing the last percent of effi- ciency from a power supply generally place a Schottky diode in parallel with the synchronous rectifier MOSFET.

    This diode conducts only during the dead time. A Schottky diode in parallel with the silicon body diode turns on at a lower voltage, ensuring that the body diode never conducts. Generally, a Schottky diode used in this way can be smaller and cheaper than the type the simple buck circuit requires, because the average diode current is low. Schottky diodes usually have peak current ratings much greater than their DC current ratings. Conduction losses during the dead time can become significant at high switching frequencies. Light-load efficiency is a key parameter for mobile applications in which the computer spends a long time in a nearly dormant suspend mode.

    When load current is light, the inductor current discharges to zero, becoming discontinuous or reversing direction.

    DSP-Based Electromechanical Motion Control (Power Electronics and Applications Series)

    Many control strategies have been proposed to deal with this problem. The fundamental prin- ciples of controlling the synchronous switch are briefly described in the preceding section. The param- eters are kept exactly the same for the synchronous buck converter. This is necessary to check the efficiency improvement by synchronous rectification. This loss is 2. This figure confirms that there is no difference in operation compared with conventional buck converter.

    For an 18 W application, this loss results in Higher battery voltage and lighter load current enhance the value of a synchro- I Lo Vo 8. I Lo Vo The duty ratio for the main switch increases with the bat- tery voltage.

    Uh-oh, it looks like your Internet Explorer is out of date. For a better shopping experience, please upgrade now. Javascript is not enabled in your browser. Enabling JavaScript in your browser will allow you to experience all the features of our site. Learn how to enable JavaScript on your browser. Although the programming and use of a Digital Signal Processor DSP may not be the most complex process, utilizing DSPs in applications such as motor control can be extremely challenging for the first-time user.

    DSP-Based Electromechanical Motion Control provides a general application guide for students and engineers who want to implement DSP-based motion control systems in products and industrial systems. This overview explains the benefits of integrating DSP into motion control, detailing the degree of freedom provided by a a DSP for the development of constructive, computationally extensive algorithms. The authors explain how the use of these advanced algorithms can drastically increase the performance and efficiency of an electromechanical system. Chapters are supported by laboratory exercises, enabling you to immediately apply the information to practical scenarios.

    Following an extensive analysis of the LF DSP processor, the book presents numerous real-world applications, demonstrating current use and inspiring future development. Park and Clarke's Transformations. Space Vector PWM.